1. Field of the Invention
The present invention relates to motor control systems, and more particularly to a spin motor control system for disk drive apparatus.
2. Description of the Relevant Art
Magnetic disk storage systems such as hard disk drive systems and floppy disk drive systems have been and continue to be the predominant mechanisms for providing large volumes of relatively low-cost computer accessible memory or storage. A typical hard disk drive system includes a number of adjacently positioned disks coated with an appropriate magnetic material that are mounted for rotation on a common spindle. The typical system further includes a set of transducer heads carried in pairs on elongated supports for insertion between adjacent disks wherein the heads of each pair face in opposite directions to engage opposite surfaces of the adjacent disks. The transducer heads transform magnetic variations into electric variations when reading data stored on the disks, and transform electric variations to magnetic variations when writing data to be stored on the disks. The support structure is coupled to a positioner motor that typically includes a coil mounted within a magnetic field for linear movement and is typically oriented relative to the disks to move the heads radially over the disk surfaces to thereby enable the heads to be positioned over any annular track on the surfaces. During normal operation, the positioner motor, in response to control signals from a host computer, positions the transducer heads radially for recording data signals on, or retrieving data signals from, a pre-selected one of a set of concentric storage tracks on the disks.
A typical hard disk drive system also includes a spin motor operatively connected to the spindle for rotating the magnetic disks during data read and data write operations. An electronic control and driving circuit is coupled between the spin motor and the host microprocessor interface to provide drive signals to the motor windings to thereby control the speed and other operating parameters of the spin motor, as well as to control initial start-up of the spin motor.
FIG. 1 is a schematic view of a portion of a three-phase brushless spin motor 10 connected to associated control and drive circuitry. For the particular example illustrated herein, spin motor 10 is a twelve pole motor having nine windings. The nine windings are grouped into three sets, wherein each winding set is selectively driven at a predetermined phase and is represented by one of phase windings 12, 14, 16. As known to those skilled in the art, a sequencer 18 and a motor amplifier 20 collectively operate to selectively drive the phase windings 12, 14, 16 in a manner as explained below to thereby induce rotation of the rotor shaft of motor 10.
Referring next to FIG. 2A in conjunction with FIG. 1, traces 1, 2, and 3 illustrate the motor torque generated when a constant current flows through selected pair combinations of phase windings 12, 14, 16 with respect to motor electrical degrees. Trace 1 shows the motor torque curve with respect to electrical degrees when transistors 20a and 20f are turned on (20b-20e turned off), resulting in the flow of current from phase A winding 12 to phase C winding 16. Similarly, trace 2 shows the motor torque curve on the common horizontal axis when transistors 20a and 20d are turned on, resulting in the flow of current from phase A winding 12 to phase B winding 14. Finally, trace 3 shows the motor torque curve when transistors 20d and 20e are turned on, resulting in the flow of current from phase C winding 16 to phase B winding 14. The extremum torque points occur 60 electrical degrees apart. For a twelve pole motor, 360 electrical degrees correspond equivalently to 1/6 of a mechanical revolution of the rotor.
To spin the rotor in one continuous direction, the motor torque must be either continuously positive or continuously negative. A continuously positive motor torque, for example, can be provided by designing and controlling sequencer 18 to turn on selected pairs of the transistors 20a-20f in a predetermined and precisely timed sequence to thus result in an overall torque curve as defined along the extremum segments connecting a1-a7. It should be noted that the curve connecting segments a4-a7 results from the flow of current in a reverse direction through the respective phase winding pairs. The overall torque curve defined along the segments a1-a7 results in maximum torque with the least ripple, and thus is considered the result of optimal commutation timing. The predetermined sequence required for turning on the transistors 20a-20f as controlled by sequencer 18 is as follows:
Sequence 1: Transistors 20a and 20f turned on - - - Current flows from phase A winding 12 to phase C winding 16 - - - Generates torque segment a1 to a2; PA1 Sequence 2: Transistors 20a and 20d turned on - - - Current flows from phase A winding 12 to phase B winding 14 - - - Generates torque segment a2 to a3; PA1 Sequence 3: Transistors 20d and 20e turned on - - - Current flows from phase C winding 16 to phase B winding 14 - - - Generates torque segment a3-a4; PA1 Sequence 4: Transistors 20b and 20e turned on - - - Current flows from phase C winding 16 to phase A winding 12 - - - Generates torque segment a4-a5; PA1 Sequence 5: Transistors 20b and 20c turned on - - - Current flows from phase B winding 14 to phase A winding 12 - - - Generates torque segment a5-a6; and PA1 Sequence 6: Transistors 20c and 20f turned on - - - Current flows from phase B winding 14 to phase C winding 16 - - - Generates torque segment a6-a7.
The control system for triggering the sequencer 18 typically includes a circuit for generating triggering pulses while the motor is spinning, a startup circuit for generating triggering pulses to initially spinup the motor from a stalled condition, and a monitor circuit for detecting and correcting the direction of rotation and for providing a "blanking" signal as will become evident when the following description is fully appreciated. Each of these circuits is considered separately below.
In early hard disk drive systems, the commutation timing of the brushless motor as it was spinning was controlled using Hall Effect sensors which were placed within the motor. As disk drives shrunk in size (31/2 and 21/2 inch form factors), space became extremely limited and thus the Hall sensors were removed from the spin motor to decrease its size.
In accordance, another method was developed to determine the optimal timing required for triggering the sequencer circuit to thus commutate the spin motor. This method involves the phenomenon of back electromotive force (BEMF). The BEMF signals generated for a three phase motor when measured with respect to the center tap are shown as signals 4, 5 and 6 in FIG. 2B. It is evident that the BEMF signals cross the zero voltage axis when the motor torques are at their extremum values. To provide the least amount of torque ripple, the motor is commutated at 30 electrical degrees before and after the extremum torque points. These ideal motor commutation times are shown both in FIG. 2C and in FIG. 3A.
Analog comparators are connected across each phase winding 12, 14, 16 of the motor to determine when each of the BEMF signals is greater than zero. The output signals generated by these comparators are shown in FIGS. 3B-3D. The comparator signals of FIGS. 3B-3D are logically decoded to generate the tachometer signal as shown in FIG. 3E. Such generation of the tachometer signal is known to those skilled in the art. It is noted that the optimal motor commutation times are shown to occur at the midpoint of each high and low state of the tachometer signal as represented at points X and Y, respectively.
The midpoints X and Y of each high and low state of the tachometer signal are determined in accordance with the circuits of FIGS. 4A and 4B. The voltage waveforms generated across the capacitors 22 and 28 of the circuits are shown in FIGS. 3F and 3G, respectively. To generate the waveform of FIG. 3F, capacitor 22 is charged with a constant current source 24 during the high period of the tachometer signal and is then discharged at twice the rate with a constant current sink 26 where the tachometer signal changes states. When the spin motor is running at nominal speed, the capacitor 22 reaches its lowest level at point Y which is the desired time to commutate the motor. Capacitor 22 is combined with additional sensing and triggering circuitry connected to the sequencer 18 to thereby commutate the motor amplifier 20 to the next phase.
The capacitor 28 of FIG. 4B is provided to determine the commutation points labeled X. This is accomplished by charging the capacitor 28 with a constant current source 27 during the time at which capacitor 22 is being discharged, and then holding the voltage charged until the tachometer signal changes to a high state. At this time, the capacitor 28 is discharged with constant current sink 29 that has the same magnitude as but the opposite polarity of current source 27. When the capacitor 28 reaches its minimum voltage level, sensing and triggering circuitry connected thereto senses the minimum voltage condition and thereby causes sequencer 18 and thus the motor amplifier 20 to toggle to the next phase state of the sequence.
The above-described BEMF technique for determining commutation timing works well in that when the spin motor is first starting up, the commutation points are not fixed in time. If the values of capacitors 22 and 28 are chosen correctly, the method can be used to commutate the motor even during the initial spinup of the motor. During the initial spinup of the motor, the frequency of the tachometer signal varies. The upper charge levels of capacitors 22 and 28 is not critical, and thus if the period is longer, the capacitors 22 and 28 will simply charge to a higher level. When the tachometer signal changes states due to a zero-crossing of the BEMF signal, the respective capacitor 22, 28 will be discharged. When the lowest voltage level or some other predetermined voltage threshold level is reached, the sensing and triggering electronics sequences the motor to the next commutation state.
Although the technique is seemingly ideal in principle, several disadvantages are associated therewith. Firstly, several factors associated with current sources 24, 27 and with the current sinks 26, 29 are critical. Current sources 24, 27 and current sinks 26, 29 must be well matched. If they are not precisely matched, the commutation points are incorrectly determined.
In addition, the absolute values of the current sources 24, 27 and sinks 26, 29 must be well controlled from one spindle driver integrated circuit chip to another. If the absolute values are not well matched, the dynamic range of speed control becomes inconsistent from one unit to the next.
Furthermore, since the current sources 24, 27 and sinks 26, 29 are usually in the low microamp range, they are subject to various leakage paths on a printed circuit board. This problem can effect the time constant of the respective capacitor 22, 28 being charged and discharged and thus can cause the motor to be commutated at a non-optimal time.
Several factors associated with capacitors 22, 28 are also critical. Capacitors 22, 28 may change in capacitance value due to temperature and humidity conditions. This can also be detrimental to the dynamic operating range of the speed control system.
In addition, the type of dielectric used for the capacitors 22, 28 must be considered. Material such as X7R exhibits a piezoelectric effect which causes a noise pulse at the end of the respective capacitor 22, 28 discharge cycle which can cause the sensing electronics to cause commutation of the motor to the next phase at the wrong time.
Furthermore, the values of capacitors 22, 28 must be relatively large, and thus it becomes impractical to place them within the spindle driver integrated circuit chip.
Finally, if the timing requirements for commutation of a particular system must be modified, the capacitors 22, 28 must typically be changed which therefore involves rework of the printed circuit board.
The above description considers the commutation of the spin motor phase windings 12, 14, 16 during normal operation when the motor is spinning. The commutation of the phase windings 12, 14, 16 is next considered at initial operation when the spin motor is started from a stalled condition. When the motor is stalled, there is no generated BEMF signal, and the motor must be spinning at a certain speed in order to generate an adequate BEMF signal to drive the above described sensing electronics to control commutation.
Thus, from a stalled condition, the motor is typically treated as a step motor and is thereby caused to rotate at a constant speed. This speed is determined by the parameters of inertia, torque constant, number of poles, and current applied.
One known implementation for spinning up the spin motor involves the use of a capacitor 30, a current source 32, a pair of current sinks 34, 36, an electronic switch 38, comparators 40, 42, and a pair of oneshot circuits 44, 46 as illustrated in the schematic of FIG. 5. Other components described earlier are also included in the schematic and are numbered similarly. Referring to the schematic in conjunction with the waveforms of FIGS. 6A-6C, during initial spinup when the motor is spinning too slowly to generate a sufficient BEMF signal, switch 38 is in position 1 and capacitor 30 is thereby charged by current source 32. The voltage charged across capacitor 30 is shown as segment A-B in FIG. 6A. When the voltage across capacitor 30 reaches the reference voltage VR1, comparator 40 provides a control signal to a control logic circuit 48 that responsively causes switch 38 to move to a position 2. When switch 38 is moved to position 2, current sink 34 is connected to discharge capacitor 30 until a voltage equal to voltage reference VR2 is reached. When comparator 42 senses a voltage less than reference voltage VR2, the control logic toggles switch 38 back to position 1. The capacitor 30 voltage thereafter starts to charge again along the segment C-D. Comparator 42 also generates a positive pulse (when the voltage across capacitor 30 exceeds reference voltage VR2). The pulse is received at oneshot circuit 44 which responsively generates a commutation triggering pulse to the sequencer 18. The sequencer 18 controls the phases of the motor amplifier that are active. This process repeats until the motor gains sufficient speed to start generating the BEMF signal.
When point H is reached, a BEMF commutation pulse is generated. This pulse triggers oneshot circuit 46 which thereby causes the control logic 48 to toggle switch 38 to position 3. Capacitor 30 discharges for a given period of time, as shown following point H. This discharge time is determined by control logic 48. As the number of BEMF pulses generated increases, the voltage across capacitor 30 is repetitively discharged by current sink 36 and is discharged below voltage reference VR2. When this occurs, the output signals from the two comparators 40, 42 remain low without transitions and thus the sequencer 18 no longer triggered by oneshot circuit 44. The sequencer 18 is thereafter triggered solely by the BEMF pulse through oneshot circuit 46.
FIG. 6B illustrates the commutation pulses generated by the startup oscillator oneshot circuit 44 and FIG. 6C illustrates the commutation pulses generated by the BEMF oneshot circuit 46. The faster the motor rotates, the more numerous the BEMF pulses. These rapidly occurring BEMF pulses result in the voltage across capacitor 30 to be discharged below VR2 and thus disables the startup oscillator.
Several disadvantages are associated with this type of startup oscillator. The value of capacitor 30 must be changed if the system parameters are changed. In addition, capacitor 30 is an external part to the spindle driver chip and thus requires valuable room in small form factor disk drives. Finally, current leakage from the capacitor 30 to ground can effect the startup commutation frequency. This can cause the spin motor to fail to reach nominal speed in the allotted time. Small drives typically require a fast spinup time characteristic, and hence if the startup capacitor 30 has excessive leakage, the drive will fail the requirement.
Another method used to start motors involves a variable-time timing circuit used to step the sequencer. The time characteristics for the timer is based upon the motor and the load parameters. This technique must be very conservative since it is configured in an open loop orientation. If the timing characteristics are too aggressive, the motor will fail to spinup. With conservative timing characteristics, the motor spinup time is relatively long which is a drawback for use in small disk drives. The conservative timing characteristics are in part required to accommodate changes in load, motor parameters and environmental conditions.
Another important aspect to be considered in the design of a spin motor and the associated control circuitry involves a monitor circuit. The monitor circuit for a brushless DC (BDC) motor serves two functions. The first function is to degate or blankout momentarily the BEMF commutation circuitry whenever a motor phase winding is turned off. Referring to the schematic of FIG. 7 and the associated waveforms of FIGS. 8A-8C, consider first a current flowing from phase A winding 12 to phase C winding 16. This current causes the rotor to rotate from point a1 to point a2. When point a2 is reached, the sequencer 18 commutates the motor 10 such that current flows from phase A winding 12 to phase B winding 14. Phase C winding 16 is the phase winding that is now used to measure the BEMF signal since it is the turned-off winding. However, before the BEMF signal can be monitored, the residual current in the winding 16 due to the prior phase must be allowed to discharge through a diode 60 to a storage capacitor 62. The time required to discharge the phase winding is referred to as blanking time, diode flyback, or reverse recovery time. The BEMF signals generated by each phase winding are shown in FIG. 8B. The noise "glitches" shown in each BEMF signal 4-6 result from the residual current discharge explained above and cause the signals to cross through the zero voltage level, for example, at points W and Z. FIGS. 9B, 9C, and 9D are waveforms of the BEMF comparator output signals for each phase winding and indicate when each phase winding has a positive BEMF signal. The noise glitches of the BEMF signals 4-6 cause corresponding glitches (labelled as points N on the traces) in the comparator output signals which thus interfere with the generated tachometer signal. For this reason, a delay time shown as segment D is required following a commutation triggering pulse before the BEMF signal is processed. Thus, the first task of the monitor circuit is to provide the delay after a commutation pulse.
The second function of the monitor circuit is to detect if the motor is rotating in the proper direction. Referring back to FIGS. 8A and 8B, if phase A to C is energized at point a1 and the motor is rotated to point a2, the BEMF signal 6 of phase C winding 16 should have a negative polarity (segment E) immediately following the commutation trigger pulse and delay time (segment D) described above. If the signal has a positive polarity, the motor is spinning in the wrong direction, and therefore the motor should be commutated to the next phase. This allows the motor to catch up and start generating torque of the proper polarity.
The two functions of the monitor circuit have traditionally been accomplished using current sources and comparators. FIG. 10 shows a schematic of a monitor circuit for providing a blanking delay and for providing false BEMF detection and direction correction, and FIGS. 11A-11E show the waveforms generated. One of three actions trigger the circuit: a BEMF commutation pulse provided at line 80, a startup oscillator commutation pulse provided at line 81, or the internally generated monitor false polarity correction pulse provided at line 82. A trigger pulse received at any of lines 80, 81, 82 is provided to control block 84 through OR gate 85. When a trigger pulse is received, control block 84 causes switch 86 to close and thereby causes charging of capacitor 87 with current source 83. In addition, control block 84 causes flip-flop 88 to set to a high state. The time period from the occurrence of the trigger pulse until capacitor 87 charges to voltage reference VR3 is the delay time (segment D) as shown in FIGS. 11A and 11B. When the delay pulse of FIG. 11B is high, the BEMF comparators are degated and are thus not allowed to change states. When the voltage across capacitor 87 reaches voltage reference VR3, comparator 93 enables AND gate 89. Comparator 90 monitors (at line 99) whether the polarity of the BEMF signal from phase C winding 16 is negative while AND gate 89 is active (during the time period when the pulse as shown in the waveform of FIG. 11C is high). If the BEMF is negative, OR gate 91 provides a signal to control block 94 which thereby causes capacitor 87 to be discharged by closing switch 95 (FIG. 11D). OR gate 91 also provides a signal that causes flip-flop 88 to be reset to a low state.
If the BEMF signal from phase C winding 16 is positive, then the output of AND gate 89 remains inactive. Capacitor 87 thus charges to voltage reference VR4 at point F (FIG. 11E) and thereby generates a false polarity commutation pulse as shown in FIG. 11F by means of comparator 96. The false polarity commutation FPC pulse signal is received at control block 94 through OR gate 91 that accordingly discharges capacitor 87. The FPC pulse signal is also received at control block 84 through OR gate 85 such that when capacitor 87 has been discharged, it will be recharged by current source 83 and the process repeated. The FPC pulse signal is finally provided to the sequencer so that the motor is commutated to the next state.
The blanking and direction control circuit described above has disadvantages in that an external capacitor 87 is required. In addition, the circuit does not optimally accommodate for changing motor parameters and environmental conditions.
Numerous other magnetic disk storage systems and components thereof relating particularly to spin motor control have been disclosed. Of general interest in the field of spin motor control are U.S. Pat. No. 4,933,785 to Morehouse et al., issued Jun. 12, 1990; U.S. Pat. No. 4,568,988 to McGinlay et al., issued Feb. 4, 1986; U.S. Pat. No. 4,638,383 to McGinlay et al., issued Jan. 20, 1987; U.S. Pat. No. 4,371,903 to Lewis, issued Feb. 1, 1983; U.S. Pat. No. 4,737,867 to Ishikawa et al., issued Apr. 12, 1988; and the publication "Quantum Low Power Products: Go Drive-21/2-inch Hard Disk Drives-ProDrive Gem Series-31/2-inch Small Frame Devices-Technical Highlights", September 1990.